Designing a 20/40 band CW rig – Part 4

Hi all,

This is a continuation of an earlier post, which can be found here.

Work has been continuing on my CW rig design, and now I will share another subsystem of the rig. I will get to the audio, but in this post I will be looking at the mixing function.

The SA612

Many rig designs use a SA612 mixer oscillator IC. This device is a double balanced mixer and oscillator. It can actually be used with either balanced or unbalanced RF inputs, and a balanced or unbalanced output can be taken. Regarding the oscillator, essentially only the tank circuit is external. The SA612 can also be driven by an external LO input.

Why not use it? Good question. My early concepts made use of it, but the limitation of the SA612 is its dynamic range. The third order Intercept (IP3) is at about -13dBm with a -45dBm signal. The 1dB compression point is -20dB, but given the IP3 at -13dB, a signal at this level of power would be full of inter-modulation distortion. I wanted something that would give much better inter-modulation distortion performance. In looking around, I found nothing that was quite like the SA612, which is why so many designs make use of it.

You’ll be able to see what I have done in the circuit below. This is a screen shot, with the RX signal coming in from the top from the TX blocking circuit discussed in previous posts. Note you can click on the picture for a zoomed in view:

CW rig mixing circuit

The minicircuits ADE-1L

Minicircuits have a large variety of mixers available. Most service microwave requirements, but some work down to HF. I had a good look at a number, but the ADE-1L took my interest. This device has as 1db compression at 0dBm, which is 20dB better than the SA612, along with a IP3 at 17dBm, 30dB better than the SA612. Dynamic range is therefore effectively 20dB better, and inter-modulation distortion performance should be 30dB better. Using these devices should make for a much better rig. Unfortunatley, there will need to be more work to use one (or two) of these than SA612 ICs. Lets get into dealing with the issues.

50 ohm input and output

The ADE-1L is designed for 50 ohms. The SA612 has a high input impedance, 2k is ok to feed it, while it has an output impedance of 1700 ohms. 50 ohms sounds better, but it is harder to use. What is coming in might be expected to be 50 ohms at the antenna, but by the time it gets through the TX blocking circuit, it is around 10 ohms. Essentially both devices need matching circuits – which is band specific. I discussed these networks more on a previous post.

The first mixer

The the first ADE-1L mixer, M1, takes the RF input and a LO. The RF is 7 to to 7.3MHz or 14 to 14.3MHz depending on the band. If 7 is mixed with 11, 4 is output. If 14 is mixed with 10, then 4 is output. This LO needs to change with tuning, needs to be around 3dBm, and these things can be done by the AD9834. I will look at this device more when I look at the microprocessor control, but it can be used to generate the carrier at 7 or 14 (and up the band) during TX, and generate the around 10 or 11 MHz during RX for mixing.

With these two signals mixed, I have output at 4MHz. I also have an image frequency at 18MHz for 40 metres, and 24MHz for 20 metres. There might be other products coming out of the mixer as well, but these will be well down on the 4MHz and the image frequency. I should not have much RF and LO coming out, given the isolation performance of the mixer.

Some amplification

All I have done with the RX coming in so far is to impedance match it to the output finals (which are like leaky closed switches during RX) pass through the TX block and then match it to 50 ohms for the mixer. The mixer loses 5.5dB mixing, and there would be expected to be 1 or 2 dB across all of the matching networks. There is a little over 1dB across the TX block. All of this is not such a big deal because we are dealing with 20 and 40 metres, and we don’t need to worry about noise performance so far. The SA612 actually amplifies the output as well as mixing it. There is around 5dB of loss in mixing, and then 22dB of gain, leaving a net 15dB gain with over 5dB noise figure (because of the mixer).

I have put a 4401 device to do this amplification, because the ADE-1L is only a mixer. The 50ohm output of the mixer is impedance matched to 700 ohms. This allows a lower current biasing network on the transistor. A BJT is used to keep things linear as they do this job better than MOS devices, especially at these low current levels. The 4401 is designed to give 23dB gain. I use it to do a small impedance transformation, back down to 400 ohms, a suitable level for the crystal filter about to come.

The crystal filter

I have an intermediate frequency of 4MHz with this design. Now using a series of 4MHz crystals, I can have a narrow pass band filter. The great thing about crystals is that they can be used with some series and shunt capacitors to give various types of high performance filters, such as Butterworth or Tchebycheff filters. Butterworth are a little lower performance, but have no ripple. I have designed this for a small amount of ripple (1dB) with steeper skirts, making it a Tchebycheff. The 330pF and 390pF capacitor radio decides the ripple. The amount of capacitance overall decides the pass bandwidth. I have designed for 500Hz. It is a reasonable compromise between selectivity and usability. A SSB filter width is too much for a CW rig. I think 300Hz is too tight, except in contests, but I am not really designing it for contests, more for SOTA activations.

The number of crystals forms the number of “poles”. 4 poles seems to be a good compromise for these filters, and many CW rigs have settled on this number. I will too. After the 4 crystals, I have a matching network to bring the impedance back down to 50 ohms for the second mixer.

The second local oscillator

The second mixer needs another local oscillator, this time pulled just off 4MHz, so that I get audio out. If I was using a SA612, I just need a tank circuit, but here I need something more. I need an oscillator, plus I need to get the LO up to 3dBm for the ADE-1L.

I spent quite a bit of time on this circuit. One of the problems is that the crystal can make the input to the first transistor quite free of harmonics, but the output is not much so. I found that it worked better with quite a high impedance biasing network. I also have gone with a Clapp oscillator fed from the emitter of the active device. The output has a collector resistor, but no inductor. This allows a moderate impedance path for the harmonics to go. The desired output goes through a series resonant circuit, to pass the fundamental, and then a parallel resonant circuit to shunt any remaining harmonics to ground. Most of the harmonics leave through the collector. The approach works quite well. I then use a second active device to bring the oscillation up to 3dB and impedance match to 50 ohms. Again, I have no inductor on the collector, so any harmonics (there is still a small amount) are shunted to AC ground.

At the end of all that, I have a near 4MHz local oscillator, controllable through a varicap on the crystal, mixing with a 4MHz intermediate frequency. This will yeild audio frequency output.

There will be, of course another 5.5db loss across this second mixer, so I have a net -5.5 + 22 – 5.5 for 11dB gain across all the mixers. S9 has gone from about -70dBm to 59dBm.

Next up is the audio circuit, which includes a automatic gain control. I’ll look into that for the next post.

Regards, 73, Wayne VK3WAM

Designing a 20/40 band CW rig – Part 3

Hi all,

This is a continuation of Designing a 20/40 band CW rig – Part 2.

Too many harmonics

After the last post, that looked at the TX circuitry and filters, I went back and looked at the TX efficiency. I was also a little unhappy that there was quite a bit of power in the harmonics, which were being shunted to ground. One effect of all of this was the BS170s were presenting a variable input impedance to the previous BJT driver stage. Now both MOSFETS and BJTs are current sources, the former voltage controlled and the latter current. The voltage that is present on the output of an active device is not generated by that device directly, rather the device generates current. The voltage is a function of that current and the load. Now if the load is variable, then the voltage is going to change, even if the current output from a device is a nice clean sine wave, the voltage will not be so if the load is changing. We could kind of get away with this if the load device is a BJT as it is current controlled, but MOSFETS are voltage controlled.

The answer is not more filtering, the answer was to go back and look at the design. I first tried to lower the resistances in the BS170 bias network to swamp out the impedance changes. This worked to some degree, but it was not a final solution:

  1. More current in the bias network is overall power efficiency loss. We want to keep bias currents lean.
  2. My driving device, a MMBR941 has a maximum mean collector current of 50mA and I was within 20% of this. In the words of a well known Star Trek character “I cannot give her anymore power capt’n”.

Replace the MMBR941 with another device

It takes a long time to search around for suitable devices, and they also need to not be expensive. Instead of a long search, BS170 devices are nice and cheap, so I began with using another one of these to drive the 3 BS170 finals. It was easy enough, I already have a bias divider network to set the bias, and by moving this up and down, I can control the drain current. All of the stuff on the collector and emittor of the BJT can go, except for what is now the drain inductor.

This approach gave me more drive, and allowed greater 2nd network bias currents, but the additional losses here about matched the gains post BS170 finals. Not too good.

True Class C design

The general idea of Class C is that the active device is operating for less than 50% of the cycle. This is true, but a tank circuit is needed as well. My initial design had one of these, but on the output. The literature that I saw has the tank network between the voltage supply rail and the drain. So I needed to change what is basically a resonant low pass filter on the output to a tank at the traditional spot, between supply and the BS170 drains.

This approach certainly worked, and the output was much cleaner, but not near perfect. One thing that is needed is that the resonant tank circuit needs very low values of L and high values of C. Being a shunt, it is still presenting a very low impedance to the finals on harmonic frequencies. Because the fundamental load frequency also has a lowish impedance of around 10 to 12 ohms, the shunt network needs to have a net reactance well below this on the harmonics. A bit tough to do, and still we have the problem of variable impedance being presented to the driver stage.

Class E design

I decided to go for an alternative approach, a Class E design. The circuit I showed in the previous post has a bit of a change:

CW dual band 40m 20m transceiver with Class E finals

As can be seen on the circuit, a BS170 is being used as a driver. I have also gone from three BS170 finals to two.

A Class E amplifier uses a shunt capacitor across the transistor to complete the waveform, along with a series resonant circuit. This presents high impedance to harmonics, while passing the desired frequency. Actually, one of the tricks with Class E is that this resonant circuit is not centred on the frequency of interest, but a little below. A calculator is available here, thanks to Alan G3NYK. That site has a LF orientation, but the principle certainly works on HF or higher.

Class E amplifiers also have a reputation for higher efficiency than Class C, and I am finding this to be the case. I was almost able to get 5 watts RF output from a single BS170 on 20m, and could do it easily on 40m. We want 5W on both bands, with a bit of margin on the devices, so I am still using 2. By designing for a specific Q on the resonant circuit, and the selection of the shunt capacitor, various power levels are achievable for a given input to the transistors.

With the change of circuit design, I also had to review how I could dampen the oscillator input when not TXing. Unfortunately, I have found that the leakage currents on the various MOSFET devices I have used are too great to effectively shut it out. I also tried using a NDT2955 device to cut the power to the finals when not TXing. This device has a very low voltage drop across it when used as a switch, even at several amps. Unfortunately it sets up some strange oscillation in the Class E circuit, so no success there. It really didn’t make much difference, because the BS170 will act the same way on a small signal, allowing it to leak through, regardless of the gate bias set well below cutoff, or the power being cut from the drain. In the end, it was back to a simple 4401 to sink away the unwanted signal when not TXing, and this had the best effect between the driver and the finals.

What was the effect of all of this: I achieved a efficiency on the finals of 81% on 20m and 85% on 40m. I had to tone down the circuit on 40m, because the circuit adapted directly from 20m at 5 watts produced 7.5 watts on 40m. I also would not want to change the biasing networks based on the band, so the flexibility of being able to control power gain in the components of the Class E amplifier itself was nice.

I also was able to significantly reduce the current on the final bias network. The load impedance presented to the driver from the finals is still quite variable, but with Class E, it does not matter as the output is remarkably clean. Some Class E approaches are fed directly with a square wave and this can allow for even greater efficiency. I could look into this, but I think I have captured all the low hanging fruit. I still had to allow a moderate level of current on the first bias network, but it is only about 2mA. There is about 40mA spent on the driver and less than 1mA on the second bias network. This is a total of less than 45mA, for around 550mW. In total, including the driver and bias networks, we should be spending less than 7W to drive 5W of RF on 20m and less than 6.5W on 40m. These levels are well below half of a FT-817 (not counting the FT-817’s draw for other things like the screen, activating the coil on the rear connector, etc).

The RX circuit needed some adjustments to deal with with the change in the output signal. I adjusted the bias on what is now Q12 up to about 11V, as the peak to peak voltage of the TX output here. I also need the base of the signal at least around 0V, otherwise the body diode of Q12 will begin to conduct. We will still be well within the breakdown limits of the device. Both the BS170 and 2N7002 have 60V drain source breakdown

Side note: What is this body diode on the 2N7002? It is what comes with the territory with MOSFETS, it is part of their nature, and for a N channel, it means that the drain will get passed through to the source if the drain is below the source, like a forward biased diode. Also, when reversed biased – the normal usage of a MOSFET, the diode acts like a zener diode, with the drain to source breakdown voltage being the zener voltage.

Speaking of zener diodes, I will be putting one across the BS170 finals, at somewhere in the high 40V, I’ll just need to select the device. Of course we never want to see the MOSFET put to its breakdown voltage. One impact of Class E design is that there are higher voltages than what would be possible with Class A or even Class C.

The Class E matching reactive components are of course band specific, so the location of the relay will need to change, with it being right up on the BS170’s. The peak current across the relay is about 1A, or 700mA RMS. This is well within the limits of the G6SK-2F unit I plan to use.

Latching relays

One other thing I should mention is that I am using two coil latching relays. These will only use power when I need to change the relay orientation. I’ll use 5V relays with the supply fed by a 5V regulator.

Next time, I’ll look at the audio part of the circuit.

73 de Wayne VK3WAM

Designing a 20/40 band CW rig – Part 2

Hi all,

This is a continuation of Designing a 20/40 band CW rig – Part 1.

Shown below is a schematic of the TX and initial RX parts of the rig. I have been successful in simulating all of these parts, and have a reasonable amount of flexibility if the real world performance of the components does not match the simulated performance.

Schematic of the TX and initial receive sections of the rig

TX Driving

This being a CW orientated rig, all that is needed is a oscillation on the desired TX frequency. CW merely turns it on and off. I did mention that I wanted to retain cabability of PSK and FSK modes. PSK needs the phase of the oscillation to be changeable, while FSK modes need the frequency to be changeable. An Analog Device AD9834 fits the bill. Steve KD1JV is using this in his Mountain Topper. This device is not a PLL and VCO combination, rather it generates the output digitally, and then feeds a DA convertor to generate the waveform. The output is needed for two things – it is the Local Oscillator for a mixer, and can be used directly for a TX frequency. The AD9834 has a number of capabilities that I plan to use, but this will be for another post.

Unlike Steve, I have decided to take the analog output of the AD9834. This output will drive the initial mixer, and the device selected has a 50ohm load. I’ll talk about this device on another post, but I will be using a different approach to the other designs I have seen because I want more dynamic range and better inter-modulation distortion performance.

The input to my driver is going to be taken off this output. The AD9834 can provide a balanced or unbalanced output. I need unbalanced. This signal is shown as OSC. It then feeds the base of a MMBR941, Q1. This is a RF BJT device in common emittor configuration. One of the nice aspects of BJT devices in common emitter is there is a fair degree of flexibility in setting the input impedance and the output impedance of the device. I decided on a 150ohm input impedance. I am also driving the device fairly hard, but not too close to device limits. The gain on the device is 26dB.

To save power when not TXing, I use Q2 as a switch. This 4401 device is being used as a current sink, switched on during TX, and off when not. It deprives Q1 of its DC ground, so no DC current will flow through the device. The bias network is shut down, but is still kept alive on AC so that the load presented to the AD9834 is not significantly changed. The effect of Q2 kills nearly all of the output power from the collector of Q1.

TX Power

My objective is to deliver 5W into a 50ohm load (antenna). There are quite a lot of choices to go about this, but I want to keep things as simple and as cheap as possible. I considered initially using another BJT in emitter follower mode, by using Q1 to deliver the desired peak to peak voltage AC signal and then the emittor follower would supply the current. I used an inductor on the emitter to improve the efficiency, but at the end of the day, it is still being used in Class A mode. This is a CW rig, so Class C amplifiers await!

The schematic shows three BS170s that are driven by the output of Q1 on their gates. These are N-channel MOSFETs that can dissipate about 800mW of power. If they are run 66% efficient, then that means that each one can deliver 1.6 watts. The efficiency is a function of the peak to peak voltage supplied by the MMR941, and what level the bias is set at. The bias mid point needs to be below the pinch off voltage on the BS170’s for Class C operation. The further away, the better the efficiency, but it cannot be set too far away, as there are limits to the peak to peak output I can get from the driving device. After a fair amount of experimentation, I set the bias level using a voltage divider resistor network. R9 and R10 (the R10 going to ground) form this network. (The other R10 nearby on the base of Q3 has a new identity as R18). This level ends up being around 1.2V, around a volt below the BS170 cutoff to ensure Class C operation.

The job of Q3 is to act as a switch, on during TX, so that R9 and R10 do their job as described. Off during RX, so R9 is taken out of the circuit, and R10 pulls the base of the BS170’s to ground so that they are cut off from doing anything. The BS170s then act as open circuits.

Presenting the output to the load

Class C waveforms have less than half of the initial sine wave present. The BS170s also present a changing load to Q1, so what it produces is not a great looking sine wave either. We actually have quite an ugly looking waveform on the output of the BS170s. This needs to be cleaned up. Also, the load is not 50 ohms. It can’t be if this circuit is going to operate off 12V. The best that can be hoped for, in terms of an output wave form is 24V peak to peak, due to the action of the inductor L2. I used the complex model of L2 in my simulators, rather than using a perfect inductor. The complex model output was practically indistinguishable from the ideal model. Coilcraft make some good inductors, and I plan to save builders of this rig from having to wind their own inductors, by using these Coilcraft chip inductors.

24V peak to peak really only allows for 12 ohms load impedance directly on the finals. This needs to be transformed to 50 ohms at the antenna connection. All rigs need to do this transformation. Steve’s Mountain Topper Radio provides a network for each band. The FT-817 provides one for each of its bands that it can TX on, and they are switched by relays. The FT-817 set of finals are operated in a push-pull configuration, but off a 8V rail. This means that the load impedance that these are driving will be lower than 12 ohms, perhaps about 9. I should analyse the inductor/capacitor networks for a given band that Yaesu have put in there. If you were to look at the circuit diagram, these matching networks take up most of the power board module schematic.

I reckon a bit of convenience is a good thing, so I am going to use a relay to switch between the two bands. This relay is K1. It is DPDT, so I can use the one relay for both ends of the matching network.

The match network has a LC tank which does most of the job of restoring a clean sine wave, and then a LC series, presenting a high impedance to remaining harmonics. This second LC series begins the process of impedance matching, where I only need two more capacitors to complete the job. This is per band of course.

Now reality bites, and there is the need to use real world components. Also, for these networks, X7R dialectic is not acceptable, but for cheap capacitors, 1nF or above tends to be X7R. So to get around this, I have paralleled up some caps to get the values I want, and to continue to use NP0 dialectic caps in 0805 SMT size. There was one cap where this approach was too long in the tooth, so I use a 3.3nF ATC cap that is 1111 size. It’s a much more expensive cap, but I’m only using one!

After these matching components, I have a sine wave output at 50 ohms supplied to the antenna. It is the job of the operator to take it from there.


I have to receive as well, this is a transceiver after all! Now, approach 1 could take the signal straight off the antenna jack, but there is the matter of the TX output to deal with. At 50 ohms, this is not going to be a 20V peak to peak output any more. It is closer to 83V peak to peak. Of course, all of the caps need to have dialectics rated to 50V, as 83V gets 41.5V from ground, each side.

Now, I could use a relay to switch between RX and TX, to shut out the TX signal from going anywhere but out the antenna jack to the antenna. The FT-817 does. One problem – full break in on CW. It is not too good to have a relay clatter (twice) every time you send a dit or a dah. I want full break in capability on this rig, so this means using a transistor approach. A transistor is going to have to hold back this 83V, but that is too much.

Instead, I have taken a similar approach to Steve KD1JV, by putting a blocking circuit at the 12 ohm area. This cuts down the voltage that the blocking network needs to resist on TX. It also has the benefit of providing a filter through the same LC networks that dress the TX output for the antenna.

The main transistor to block is Q9, a N-channel MOSFET 2N7002. I have used a biasing network, that is assisted by BSS84 Q10 and a 4401 Q11.

Q11 is a current sink. When switched on, it takes the gates of both Q9 and Q10 to near ground. Q9 is a NMOSFET, so a ground on its gate will block any signal that is about 0V or higher. Near ground on Q10 is a PMOSFET, so it is switched on, there being nearly -12V between its gate and source. Q10 switched on takes R15 out of the circuit, leaving a voltage divider of R13 and R14, meaning Q9 is biased over 10V. The input signal is going to swing between 0V or so and 20V on the drain of Q9, with 0v at its gate. If something makes it to the source of Q9, it won’t be much below the gate, meaning Q9 should stay switched off, switching off TX signal from sensitive RX circuitry.

When wanting RX, Q11 is switched off. This then means the gates of Q9 and Q10 are pulled to 12V by R17. 12V on Q9 gate will tend to switch it on. Q10 will be switched off, meaning that the bias network is now R15+R13 against R14. The drain of Q9 will be biased at around 1V. There is 12V on the gate now, and around ground on the source. Q9 will be switched on as hard as I can make it, so get as much of the RX signal through.

Unfortunately, when TXing, quite a bit of the signal still makes it through Q9. Too much by itself to be safe. The transistor itself is ok, it is not being used anywhere near maximum limits, but the mixers awaiting later will not like what gets through. I want to block this signal, so I use another LC series resonant device to block the harmonics. This is because what does get through Q9 is horribly mangled with most of the power in the harmonics, not the fundamental. This cuts things down nicely. Also this network starts the process of impedance matching from 12 ohms back up to 50 as the mixer I want to use wants 50 ohms as a source impedance. Now this matching network is band specific, so this means a second relay. This relay, like the TX relay is only switched during band changes, so there will not be any relay clatter here. Q12 and Q13 are little helpers. They are switched on during TX to shunt away any TX signal that still makes it through this far, but because of the LC network, it ain’t much. The job of Q9 is also easier to block the TX signal when it is mostly high impedance sitting behind it.

Wrapping up

This about wraps up this post. Next up I will look at the audio processing part of the rig. The other major parts are the RX mixing and filtering and the microprocessor. I’ll look at these with later posts. My plan is to get one of these rigs built in about a month to two months as a prototype, and we’ll see how it goes from there.

Regards, 73, Wayne VK3WAM

This topic is continued at Designing a 20/40 band CW rig – Part 2.

Designing a 20/40 band CW rig – Part 1

Hi all,

I have been spending some time working on the design of a ultra portable CW rig. I’ve been spending time looking at various kit designs from Steve KD1JV, Ramsey Kits, You Kits, the Elecraft KX1 and others. Some rigs are quite simple, others have a more sophisticated approach.


In setting down to design a rig, I was wanting something that would work well in a SOTA activation context. I would want the rig to do the following:

  • I did not want to be limited to a narrow band of frequencies around a crystal, but wanted the whole band to be available.
  • The power output needed to be around 5W – full QRP. 500mW CW can be tough going in VK. One attraction with SOTA activations is working the DX chasers, so 5W is nice, while keeping the QRP endorsement intact.
  • I’d like to keep open the option of some digital modes, even if the rig would not do these directly itself.
  • The rig would need RIT.
  • If possible, to switch bands without needing to use external switches.
  • Deliver near 5W from a 12V source, safe for 12.6V – a three series LiPo cell, usable down to 9V.
  • Do all of this in a kit that would cost less than US$200, good, $150 better, $100 best – but no promises!

Overall rig design

The rig could basically be broken down into a number of sub-sections.

  1. Power – a cut-off for low input voltage, and regulators for 5V and 3.3V
  2. Audio frequency – RX audio and sidetone
  3. TX – SSB rigs mix up, but this being a CW rig, I only need to switch an oscillator output and amplify up to 5W. I also need to impedance match for 50ohms at the RF connector
  4. Oscillator – Given my tune the whole band requirement – I’m going to need a PLL and VCO or equivalent circuit, such as a DDS.
  5. RX – Taking the RF connector, I need to either use a relay to switch away the TX, or a transistor switch. I then feed a mixer to an intermediate frequency for filtering, and then can feed a second mixer to audio frequency
  6. Control – Need to use an embedded controller – I am familiar with the PIC line, so I’ll use something there. This will need to set dividers for the PLL or DDS, take button input from the user for tuning, band selection, RIT, provide a keyer facility (I won’t be using a straight key) – with the ability to configure at least the wpm send rate

I’ll go through these in greater detail over a number of posts. I have been influenced by some of Steve KD1JV’s ideas, but also want to do some of my own things. I have already mentioned that I want internal band selection. I also want a volume control, with the rig able to drive at least a 16 ohm speaker, along with low impedance head/ear phones – like the ones supplied with mobile/cell phones. I also want it to be capable of driving 200 ohm headphones.

Above, I floated ideas of how to switch TX/RX. A relay could be used, but if full break-in is going to be supported, relays are a pain. Relays are good for things like band switching – ’cause you don’t switch band 100 times a minute, but to keep the power down, they will need to be latching.

At time of writing this post, I have most of the sub-section circuits designed and thrashed through simulations and a bit of bread boarding. I’ll hopefully put up some posts over the next few days or so looking at TX, the audio section and some of the RX.

This design is going to be largely SMT, but to keep things sane, I’ll use 0805 or larger size components. I could make it thru-hole, but the physical size would be much larger, and it would be significantly more expensive.

73 de Wayne VK3WAM

Continued in Designing a 20/40 band CW rig – Part 2